Sensing apparatus and method

ABSTRACT

There is described an inductive sensor for sensing a parameter such as position. The inductive sensor includes: receive aerial which is electro-magnetically coupled to a magnetic field generator via a resonant circuit with the electromagnetic coupling varying in dependence upon the sensed parameter so that a sense signal induced in the receive aerial is representative of the sensed parameter. The magnetic field generator generates a magnetic field including a first magnetic field component at a first frequency which is operable to induce resonance in the resonant circuit and a second magnetic field component at a second frequency which is not operable to induce resonance in the resonant circuit. The signal processor processes the sense signal to generate a signal component corresponding to a first component of the sense signal at the first frequency adjusted using a second component of the sense signal at the second frequency so that a noise component of the first component of the sense signal is reduced, and determines a value representative of the sensed parameter using the signal component.

This application claims priority to International Patent Application No.PCT/GB04/000597 filed on Feb. 17, 2004, which claims priority to GBPatent Application No. 0303627.4 filed on Feb. 17, 2003 in GreatBritain.

This invention relates to a method of sensing the position or the speedof an object, and an apparatus therefor. The invention has particularrelevance to inductive sensors in which a magnetic field induces asignal in a resonant circuit.

BACKGROUND OF THE INVENTION

UK Patent Application GB 23744A describes an inductive position sensorin which a transmit aerial and a receive aerial are formed on a firstmember, and a resonant circuit having an associated resonant frequencyis formed on a second member which is movable relative to the firstmember. An excitation signal having a frequency component at or near theresonant frequency of the resonant circuit is applied to the transmitaerial resulting in the generation of a magnetic field having a magneticfield component at or near the resonant frequency of the resonantcircuit. The generated magnetic field induces a resonant signal in theresonant circuit, which in turn induces a sense signal in the receiveaerial that varies with the relative position of the first and secondmembers. The sense signal is processed to determine a valuerepresentative of the relative position of the first and second members.

In the position sensor described in GB 2374424A, the resonant signalinduced in the resonant circuit is generated as a result of anelectromotive force which is proportional to the rate of change of themagnetic field component at or near the resonant frequency. As theimpedance of the resonant circuit is substantially entirely real at theresonant frequency, the resonant signal is approximately in phase withthe electromotive force and accordingly is approximately 90° out ofphase with the frequency component of the excitation signal near theresonant frequency. The sense signal induced in the receive aerial isgenerally in phase with the resonant signal, and therefore the sensesignal is also approximately 90° out of phase with the component of theexcitation signal near the resonant frequency of the resonant circuit.

The sense signal is synchronously detected using a signal which has thesame frequency as, but is in phase quadrature with, the frequencycomponent of the excitation signal near the resonant frequency of theresonant circuit. By using such phase sensitive detection, noise whichis at the same frequency as, and is in phase with, the frequencycomponent of the excitation signal near the resonant frequency of theresonant circuit is substantially removed along with noise atfrequencies away from the resonant frequency.

A problem with such an inductive sensor is that noise can occur in thesense signal having components which have the same frequency as, but arein phase quadrature with, the component of the excitation signal nearthe resonant frequency of the resonant circuit. These noise componentsare not removed by phase sensitive detection and therefore affect theaccuracy of the position measurement. Such noise components can begenerated through signal coupling between components of the inductiveposition sensor, either directly or indirectly via a magneticallypermeable body which is in close proximity with the inductive positionsensor. This problem also arises in inductive position sensors in whicha transmit aerial on a first member is directly coupled to a receiveaerial, which includes a resonant circuit, on a second member.

SUMMARY OF THE INVENTION

According to a first aspect of the present invention, there is providedan inductive sensor for sensing a parameter, the inductive sensorcomprising a magnetic field generator operable to generate a magneticfield, an aerial electromagnetically coupled to the magnetic fieldgenerator via a resonant circuit, and a signal processor operable toprocess the sense signal induced in the aerial. The electromagneticcoupling varies with the sensed parameter so that the sense signal isindicative of the sensed parameter. The magnetic field generator isoperable to generate a magnetic field having a first oscillatingcomponent at a first frequency, which is operable to induce a resonantsignal in the resonant circuit, and a second oscillating component at asecond frequency, which is not operable to induce resonance in theresonant circuit. The signal processor is operable determine the valuerepresentative of the sensed parameter by processing the sense signal togenerate a signal component corresponding to a first component of thesense signal at the first frequency adjusted using a second component ofthe sense signal at the second frequency in order to reduce noise.

Preferably, the signal processor is operable to perform synchronousdetection of the components within the sense signal at the samefrequency as, but out of phase with, the first and second components ofthe excitation signals, and to process these components to form adetection signal from which a value representative of the sensedparameter is derived.

In an embodiment, the magnetic field generator is provided on a firstmember and the resonant circuit is provided on a second member, withrelative movement between the first and second members resulting in avariation between the electromagnetic coupling between the magneticfield generator and the aerial. In this way, the relative position ofthe first and second members is determined by analysing the sense signalinduced in the aerial.

BRIEF DESCRIPTION OF THE DRAWINGS

Various embodiments of the invention will now be described withreference to the attached Figures in which:

FIG. 1 schematically shows a perspective view of a position sensoraccording to a first embodiment of the invention,

FIG. 2A schematically shows the main components of a sensor elementwhich forms part of the position sensor illustrated in FIG. 1;

FIG. 2B schematically shows a plot of how the phase of a signal inducedin the sensor element illustrated in FIG. 2A varies with the frequencyof an applied magnetic field;

FIG. 2C schematically shows a plot of how the magnitude of a signalinduced in a sense coil forming part of the position sensor illustratedin FIG. 1 varies with the frequency of an applied magnetic field;

FIG. 3 schematically shows the main signal generating and processingcircuitry of the position sensor illustrated in FIG. 1;

FIG. 4 schematically shows in more detail the form of a quadraturesignal generator of the signal generating and processing circuitryillustrated in FIG. 3;

FIG. 5 schematically shows in more detail a coil driver of the signalgenerating and processing circuitry illustrated in FIG. 3;

FIG. 6 schematically shows in more detail a synchronous detector of thesignal generating and processing circuitry illustrated in FIG. 3;

FIG. 7 schematically shows the main signal generating and processingcircuitry of a position sensor according to a second embodiment of theinvention;

FIG. 8 schematically shows the main components of an alternative coildriver to the coil driver illustrated in FIG. 5; and

FIG. 9 schematically shows the main components of an alternativesynchronous detector to the synchronous detector illustrated in FIG. 6.

DETAILED DESCRIPTION OF THE INVENTION First Embodiment

FIG. 1 schematically shows a position sensor for detecting the positionof a sensor element 1 which is slidably mounted to a support 3 to allowlinear movement along a measurement direction (the direction x in FIG.1). A printed circuit board (PCB) 5 extends along the measurementdirection adjacent to the support 3 and has printed thereon conductivetracks which form a sine coil 7, a cosine coil 9 and a sense coil 11,each of which are connected to a control unit 13. A display 15 is alsoconnected to the control unit 13 for displaying a number representativeof the position of the sensor element 1 along the support 3.

The layout of the sine coil 7 is such that current flowing through thesine coil 7 generates a first magnetic field having a magnetic fieldcomponent B₁ perpendicular to the PCB 5 which varies along themeasurement direction according to one period of the sine function overa distance L. Similarly, the layout of the cosine coil 9 is such thatcurrent flowing through the cosine coil 9 generates a second magneticfield having a magnetic field component B₂ perpendicular to the PCB 5which varies along the measurement direction according to one period ofthe cosine function over the distance L. In particular, in thisembodiment the layout of the sine coil 7, the cosine coil 9 and thesense coil 11 on the PCB 5 is identical to the layout of thecorresponding coils of the position sensor described in GB 2374424A,whose content is hereby incorporated by reference.

The control unit 13 includes excitation signal generating circuitry (notshown in FIG. 1) for applying excitation signals to the sine coil 7 andthe cosine coil 9, and sense signal processing circuitry (not shown inFIG. 1) for processing a sense signal in the sense coil 11. In this way,the sine coil 7 and the cosine coil 9 form a transmit aerial and thesense coil 11 forms a receive aerial. In this embodiment the layout ofthe sine coil 7, the cosine coil 9 and the sense coil 11 results in theelectromotive forces directly induced in the sense coil 11 by currentflowing through the sine coil 7 and/or the cosine coil 9 generallybalance each other out. In other words, in the absence of the sensorelement 1, the sense signal directly generated in the sense coil 11 bycurrent flowing through the sine coil 7 and/or the cosine coil 9 issmall. Using the sine coil 7 and the cosine coil 9 for the transmitaerial has the further advantage that the electromagnetic emissionsresulting from current flowing through the sine coil 7 and/or the cosinecoil 9 diminish with distance at a faster rate than for a singleconductive loop. This allows larger drive signals to be used while stillsatisfying regulatory requirements for electromagnetic emissions.

As shown in FIG. 2A, the sensor element 1 includes a coil 21 whose endsare connected together via a capacitor 23. As the coil 21 has anassociated inductance, the coil 21 and the capacitor 23 together form aresonant circuit. In this embodiment, the resonant circuit has a nominalresonant frequency f_(res) of 2 MHz, although the actual resonantfrequency varies slightly with variations in environmental factors suchas temperature and humidity.

When an oscillating excitation signal having a frequency component at ornear the resonant frequency of the resonant circuit is applied to thesine coil 7 and the cosine coil 9, an oscillating resonant signal isinduced in the resonant circuit. This oscillating resonant signal varieswith the position of the sensor element 1 along the measurementdirection because the proportions of the resonant signal induced by thesine coil 7 and the cosine coil 9 vary with the position of the sensorelement 1. The oscillating resonant signal in turn induces a signal inthe sense coil 11, which will hereafter be referred to as the signalcomponent of the sense signal.

FIG. 2B shows how the phase of the signal component of the sense signalvaries with the frequency of the excitation signal. As shown, as thefrequency of the excitation signal passes through the resonant frequencyf_(res) of the resonant circuit, the phase difference between the signalcomponent of the sensed signal and the excitation signal passes from 0°to 180°, with the phase difference being 90° at the resonant frequency.

The sense signal induced in the sense coil also includes noise formed bysignal coupling between components of the position sensor, eitherdirectly (for example the signals directly induced in the sense coil 11by current flowing through the sine coil 7 and the cosine coil 9) orindirectly via a body other than the resonant circuit of the sensorelement 1. As such, the noise is a systematic error signal. Even aftersynchronous detection, this noise results in a noise component in thedetected signal. FIG. 2C schematically shows the variation in themagnitude of the sense signal after synchronous detection with thefrequency of the excitation signal. In FIG. 2C, the noise componentforms a background noise level 33 which is substantially constant overthe illustrated frequency range, and the signal component of the sensesignal forms a peak 31 above the background noise level 33 which iscentred at the resonant frequency f_(res).

According to the present invention, the excitation signal generated byexcitation signal generating circuitry has a first frequency componentat a frequency f₁ at or near the resonant frequency f_(res) of theresonant circuit and a second frequency component at a second frequencyf₂ away from the resonant frequency f_(res). The sense signal processingcircuitry synchronously detects the amplitude of components of the sensesignal at the first frequency f₁ and the second frequency f₂. Theamplitude of the component at the second frequency f₂ provides a measureof the noise level, and the sense signal processing circuitry uses themeasure of the noise level to adjust the detected component at the firstfrequency f₁ to improve the signal to noise ratio.

The excitation signal generating circuitry and the sense signalprocessing circuitry will now be described in more detail with referenceto FIG. 3. As shown in FIG. 3, the excitation signal generatingcircuitry includes a first quadrature signal generator 41 a whichgenerates an in-phase signal I₁ and a quadrature signal Q₁ at the firstfrequency f₁, which in this embodiment is 2 MHz (i.e. approximatelyequal to the nominal resonant frequency f_(res) of the resonant circuitof the sensor element 1). The excitation signal generating circuitryalso includes a second quadrature signal generator 41 b which generatesan in-phase signal I₂ and an inverted quadrature signal −Q₂ at thesecond frequency f₂, which in this embodiment is 1 MHz which issufficiently far away from the nominal resonant frequency f_(res) that asignal at the second frequency f₂ does not induce resonance in theresonant circuit.

FIG. 4 shows the main components of a quadrature signal generator 41. Asshown in FIG. 4, each quadrature signal generator 41 is formed by aconventional arrangement in which the output of a square wave oscillator63 is input to the clock input of a first D-type flip-flop 65 a, withthe inverting output of the first D-type flip-flop being connected tothe input of the first D-type flip-flop 65 a to form a divide-by-twocircuit. The output of the square wave oscillator 63 is also input, viaan inverter 67, to the clock input of a second D-type flip-flop 65 b,with the non-inverting output of the first D-type flip-flop 65 a beingconnected to the input of the second D-type flip-flop 65 b. In this way,the non-inverting output of the second D-type flip-flop outputs a signalQ which is phase quadrature with the signal I output by thenon-inverting output of the first D-type flip-flop 65 a.

Returning to FIG. 3, the excitation signal generating circuitry alsoincludes a square wave oscillator 43 which generates a modulation squarewave signal at a frequency f_(mod) of 2.5 kHz. The modulation squarewave signal is input to a pulse width modulation (PWM) type patterngenerator 45 which generates digital data streams, clocked at 2 MHz,representative of sinusoidal signals at the modulation frequencyf_(mod). In particular, the PWM type pattern generator 45 has twooutputs 46 a, 46 b with the first output 46 a outputting either a signal+SIN representative of a sine signal at f_(mod) or a signal −SINrepresentative of an inverted sine signal at f_(mod), and the secondoutput 46 b outputting a signal COS which is representative of a cosinesignal at f_(mod).

The sine signal ±SIN output by the first output 46 a of the PWM typepattern generator 45 is applied to a first digital mixer 47 a and asecond digital mixer 47 b, and the cosine signal COS output by thesecond output 46 b of the PWM type pattern generator 45 is applied to athird digital mixer 47 c and a fourth digital mixer 47 d. The firstdigital mixer 47 a and the third digital mixer 47 c mix the sine signal±SIN and the cosine signal COS respectively with the in-phase carriersignal I₁ at the first frequency f₁. Similarly, the second digital mixer47 b and the fourth digital mixer 47 d respectively mix the sine signal±SIN and the cosine COS with the in-phase carrier signal I₂ at thesecond frequency f₂. In this embodiment, each digital mixer 47 is formedby a NOR gate.

The outputs of the first digital mixer 47 a and the second digital mixer47 b are input to a first coil driver 49 a which adds and amplifies theoutputs to form a drive signal which is applied to the sine coil 7. Thedrive signal applied to the sine coil 7 therefore has a term I(t) of theform:I(t)=A sin 2πf _(mod) t(sin 2πf ₁ t+sin 2πf ₂ t)where A is a constant.

The outputs of the third digital mixer 47 c and the fourth digital mixer47 d are input to a second coil driver 49 b, which adds and amplifiesthe outputs to form a drive signal which is applied to the cosine coil9. The drive signal applied to the cosine coil 9 therefore has a termQ(t) of the form:Q(t)=A cos 2πf _(mod) t(sin 2πf ₁ t+sin 2πf ₂ t)

FIG. 5 shows the main components of each coil driver 49. As shown, thesignals output by the corresponding digital mixers 47 are input, viarespective resistors having resistance R₁, to the inverting input of anoperational amplifier 71. The non-inverting input of the operationalamplifier 71 is connected to ground, and a resistor having a resistanceR₂ is connected between the inverting input of the operational amplifier71 and the output of the operational amplifier 71 so that theoperational amplifier 71 acts as an inverting amplifier. The coil beingdriven is connected between the output of the operational amplifier 71and ground.

The magnetic fields generated by the drive signals flowing through thesine coil 7 and the cosine coil 9 induce a sense signal S(t) in thesense coil 11 of the form:

${{S(t)}{\alpha\left\lbrack {{C\;{\cos\left( {{2\;\pi\; f_{mod}t} - \frac{2\pi\; X}{L}} \right)}} + {\xi_{1}(t)}} \right\rbrack}\cos\; 2\;\pi\; f_{1}t} + {{\xi_{2}(t)}\cos\; 2\pi\; f_{2}t} + {{other}\mspace{14mu}{terms}}$where C is a constant, X is the position of the sensor element 1relative to the PCB 5 along the X-direction, ξ₁(t) and ξ₂(t) are theamplitudes of the part of the noise component in phase quadrature withthe components of the drive signals at the first frequency f₁ and thesecond frequency f₂ respectively. The other terms relate to terms atfrequencies away from f₁ and f₂ and terms at the frequencies f₁ and f₂which are in phase with the components of the drive signals at f₁ andf₂. The noise component amplitudes ξ₁(t) and ξ₂(t) at the modulationfrequency f_(mod) and frequencies which are comparatively slow withrespect to the modulation frequency f_(mod), due to changes inenvironmental factors such as movement of a nearby conductive object.

The sense signal S(t) is input to a first synchronous detector 51 atogether with the quadrature signal Q₁ at the first frequency f₁. Thefirst synchronous detector 51 a performs synchronous detection of thesense signal S(t) using the quadrature signal Q₁ as the reference signalto generate a first detection signal D₁(t) having the form:

${D_{1}(t)} = {{C\;{\cos\left( {{2\;\pi\; f_{mod}t} - \frac{2\;\pi\; X}{L}} \right)}} + {\xi_{1}(t)}}$

The sense signal S(t) is also input to a second synchronous detector 51b together with the inverse quadrature signal −Q₂ at the secondfrequency f₂. The second synchronous detector 51 b performs synchronousdetection of the sense signal S(t) using the inverse quadrature signal−Q₂ as the reference signal to generate a second detection signal D₂(t)of the form:D ₂(t)=−ξ₂(t)

FIG. 6 shows the sense coil 11 and the main components of one of thesynchronous detectors 51. As shown, a first end 81 and a second end 83of the sense coil 11 are connected to respective inputs of a switchingarrangement 85, which multiplies the sense signal by the input referencesignal (i.e. the quadrature signal Q₁ for the first synchronous detector51 a and the inverse quadrature signal −Q₂ for the second synchronousdetector 51 b). The two outputs of the switching arrangement 85 areconnected to respective inputs of a differential amplifier 87, and theoutput of the differential amplifier 87 is passed through a low passfilter 89 which removes frequency components which are above themodulation frequency f_(mod).

Returning to FIG. 3, the first detection signal D₁(t) and the seconddetection signal D₂(t) are then input to a summing amplifier 53, whichadds the first detection signal D₁(t) and second detection signal D₂(t)together. In this embodiment it is assumed, for ease of explanation,that the noise components ξ₁(t) and ξ₂(t) are so similar in magnitudethat when the first detection signal D₁(t) and the second detectionsignal D₂(t) are added together the noise components ξ₁(t) and ξ₂(t)cancel each other out. The summed signal output by the summing amplifier53 is then input to a bandpass filter 55 centred on the modulationfrequency f_(mod). The filtered signal F(t) output by the filter 55 isof the form:

${F(t)}\alpha\;{\cos\left( {{2\;\pi\; f_{mod}t} - \frac{2\;\pi\; X}{L}} \right)}$

The filtered signal F(t) is therefore an oscillating signal at themodulation frequency f_(mod) whose phase varies with the relativeposition of the sensor element 1 and the PCB 5.

The filtered signal F(t) is input to a phase detector 57 which measuresthe phase difference between the filtered signal F(t) and the squarewave modulation signal output by the square wave oscillator 43, andoutputs a signal indicative of the measured phase difference to aposition calculator 59 which calculates the position of the sensorelement 1 relative to the PCB 5 using the measured phase difference.

The modulation of the signal at the first frequency f₁ by the +SINsignal and the COS signal, any difference between the first frequency f₁and the actual resonant frequency of the resonant circuit, the low passfilter 89 of the synchronous detector 51 and the filter 55 introduce aphase shift Δ_(F) in the filtered signal F(t) which needs to becorrected for in order to obtain high accuracy position measurement. Inthis embodiment, this correction is performed by the PWM type patterngenerator 45 alternately outputting the +SIN signal and the −SIN signaland the position calculator 59 averaging the resultant measured phasedifferences, in the same manner as described in GB 2374424A.

Second Embodiment

In the first embodiment, the amplitude of the noise components ξ₁(t) andξ₂(t) at the frequencies f₁ and f₂ respectively are assumed to be equal.In practice, however, there will be a variation in the amplitude of thenoise component ξ(t) with frequency, although the variation of the noisecomponent ξ(t) with frequency is slower than the variation withfrequency of the amplitude of the signal component resulting fromresonance in the resonant circuit of the sensor element 1. Nevertheless,the arrangements described in the first embodiment do give a reductionin noise, and therefore an improvement in the accuracy of positionmeasurement.

A second embodiment will now be described with reference to FIG. 7 inwhich, in order to achieve a more accurate estimate of the noisecomponent ξ₁(t) at the first frequency f₁, the excitation signalgenerating circuitry generates components of the excitation signal attwo frequencies away from the resonant frequency f_(res), and the sensesignal processing circuitry measures the strength of the noisecomponents at these frequencies and interpolates the noise componentξ₁(t) at the first frequency f₁. In FIG. 7, components which areidentical with corresponding components of the first embodiment havebeen referenced by the same numerals and will not be described in detailagain.

As shown in FIG. 8, in this embodiment a third quadrature signalgenerator 41 c generates an in-phase signal I₃ and an inverse quadraturesignal −Q₃ at a third frequency f₃, which does not induce resonance inthe resonant circuit. In this embodiment, the third frequency f₃ is 3MHz so that the second and third frequencies are evenly spaced on eitherside of the first frequency f₁. The in-phase signal I₃ is input to afifth digital mixer 47 e together with the sine signal ±SIN output bythe first output 46 a of the PWM type pattern generator 45, and theresultant output of the fifth digital mixer 47 e is input to a firstcoil driver 111 a together with the outputs of the first digital mixer47 a and the second digital mixer 47 b. Similarly, the in-phase signalI₃ is input to a sixth digital mixer 47 f together with the cosinesignal COS output by the second output 46 b of the PWM type patterngenerator 45, and the resultant output of the sixth digital mixer 47 fis input to a second coil driver 111 b together with the outputs of thethird digital mixer 47 c and the fourth digital mixer 47 d.

Each of the first and second coil drivers 111 is similar to the coildrivers 49 of the first embodiment, as shown in FIG. 5, but with threeinput signals instead of two being directed to the non-inverting inputof the operational amplifier via respective resistors. The output of thefirst coil driver 111 a drives the sine coil 7, and the output of thesecond coil driver 111 b drives the cosine coil 9.

The sense signal S(t) induced in the sense coil 11 is input to: i) afirst synchronous detector 51 a together with the quadrature signal Q₁at the first frequency f₁; ii) a second synchronous detector 51 btogether with the inverse quadrature signal −Q₂ at the second frequencyf₂; and iii) a third synchronous detector together with the inversequadrature signal −Q₃ at the third frequency f₃. The outputs of thesecond synchronous detector 51 b and the third synchronous detector 51c, which are respectively representative of the noise component ξ₂(t) atthe second frequency and the nose component ξ₃(t) at the thirdfrequency, are input to an interpolator 153 which derives a value forthe noise component ξ₁(t) at the first frequency f₁. In this embodiment,the interpolator 153 performs a linear interpolation by averaging themagnitudes of the signals output by the second synchronous detector 51 band the third synchronous detector 51 c.

The signal output by the interpolator is input to a summing amplifier 53together with the output of the first synchronous detector 51 a, and thesense signal processing then proceeds in the same manner as the firstembodiment.

Modifications and Further Embodiments

In the second embodiment, the interpolator 153 performs a linearinterpolation of the noise component at the carrier frequency near theresonant frequency f_(res) using the noise components at two frequencieswhich do not induce resonance in the resonant circuit. It will beappreciated that the excitation signals applied to the sine coil 7 andthe cosine coil 9 could include components at more than two frequencieswhich do not induce resonance in the resonant circuit, with the noisecomponents at these frequencies being measured by respective synchronousdetectors and input to an interpolator. Further, the interpolator 153could perform the interpolation in accordance with a more complicatedfunction which more closely matches the variation of the noise componentξ(t) with frequency.

In the first embodiment, as shown in FIG. 5 the coil driver 49 includesa summing amplifier which performs an analogue summation of the twoinput signals. FIG. 8 shows an alternative coil driver to the coildriver used in the first embodiment. As shown, the two input signals areinput to a NAND gate 121 and an OR gate 123. The coil driver comprises afirst amplification circuit 125 a and a second amplification circuit 125b which are connected in parallel between the supply voltage V_(cc) andground. The first amplification circuit 125 a comprises a p-channelMOSFET switch P1 and an n-channel MOSFET switch N1 with the drain of P1connected to the drain of N1 and the gates of P1 and N1 connected toeach other. The signal output by the NAND gate 121 is input to an inputterminal located at the common gate of P1 and N1. Similarly, the secondamplification circuit 125 b is formed in an identical manner to thefirst amplification circuit 125 a using a p-channel MOSFET switch P2 andan n-channel MOSFET switch N2 and the output of the OR gate 123 isapplied to an input terminal located at the common gate of P2 and N2.The coil being driven is connected between an output terminal of thefirst amplification circuit 125 a located at the connection between thedrain of P1 and the drain of N1 and an output terminal of the secondamplification circuit 125 b located at the connection between the drainof P2 and the drain of N2.

In this way, if the signals input to the NAND gate 121 and the OR gate123 are both at the LOW level, current flows in a first directionthrough the coil being driven; if the signals input to the NAND gate 121and the OR gate 123 are both in a HIGH level, then current flows throughthe coil being driven in a second direction which is opposite to thefirst direction. If one of the signals input to the NAND gate 121 andthe OR gate 123 is in a HIGH level and the other signal is in a LOlevel, then no current flows thought the coil being driven. In this way,digital switching allows three different states of driving of the coilbeing driven and therefore summation using digital signals is possible.

In the first embodiment, separate synchronous detectors are used todetect the signal components at the frequencies f₁ and f₂. However, asingle synchronous detection operation may be performed by multiplyingthe sense signal by a reference signal having frequency components at f₁and f₂. FIG. 9 shows a synchronous detector implementing such anarrangement, together with the sense coil 11. As shown, the sense coil11 is connected to a switching arrangement 131 having two independentlycontrolled signal-pole signal-throw switches 133 a and 133 b. Each ofthe switches 133 has two input terminals connected to respective ends ofthe sense coil 11. The control signal for the switches 133 are generatedby inputting the quadrature signal Q₁ at f₁ and the inverse quadraturesignal −Q₂ to an AND gate 135 and an OR gate 137. The output of the ANDgate 135 is connected to the first switch 133 a and the output of the ORgate 137 is connected to the second switch 133 b.

The output terminals of each switch 13 is connected to a respectiveinput of a differential amplifier 139, and the output of thedifferential amplifier is input to a low pass filter 141 which removesfrequency components above the modulation frequency f_(mod).

In the described embodiments, a transmit aerial is formed by twoexcitation windings and a receive aerial is formed by a single sensorwinding. It will be appreciated that many other arrangements of transmitaerial and receive aerial in which the electromagnetic coupling betweenthe transmit aerial and the receive aerial varies along a measurementpath could be used. For example, the transmit aerial could be formed bya single excitation winding and the receive aerial could be formed by apair of sensor windings, with the respective strengths of signalsinduced in the two sensor windings being indicative of the location ofthe sensor element. In such an arrangement, the sense signal induced ineach sensor winding is adjusted using a noise component at a frequencyaway from the resonant frequency in order to reduce noise.

It will also be appreciated that the position sensor described in thefirst embodiment could be adapted to measure a linear position along acurved line, for example a circle (i.e. a rotary position sensor) byvarying the layout of the sine coil and the cosine coil in a mannerwhich would be apparent to persons skilled in the art. The positionsensor could also be used to detect speed by periodically detecting theposition of the sensor element as the sensor element moves along themeasurement path, and then calculating the rate of change of position.

As described in the first embodiment, the phase shift Δ_(F) introducedin the filtered signal F(t) is removed by effectively taking twomeasurements of the position with the phase of the signal applied to thesine coil 7 being reversed between measurements. It will be appreciatedthat in alternative embodiments, the reverse measurement need only beperformed intermittently to determine a value for the phase shift Δ_(F)which has the advantage of increasing the measurement update rate.Alternatively, a predetermined value for the phase shift Δ_(F),determined by a factory calibration, could be subtracted from a singlephase measurement. However, this is not preferred because it cannotallow for environmental factors which affect the resonant frequencyf_(res) and quality factor of the resonant circuit and therefore varythe phase shift Δ_(F).

It will be appreciated that if the phase angle measured using the −SINsignal is subtracted from, rather than added to, the phase anglemeasured using the +SIN signal then the position-dependent phase shiftwould be removed to leave a value equal to twice the phase shift Δ_(F).In an embodiment, the resonant circuit is manufactured using componentshaving a high sensitivity to environmental factors so that the variationof resonant frequency with environmental factors is the dominant causeof the phase shift Δ_(F). In this way, a measurement of the phase shiftΔ_(F) can be indicative of an environmental factor, for exampletemperature in a constant humidity environment or humidity in a constanttemperature environment. Typically, this would involve storing in thecontrol circuitry of the inductive sensor a factory calibration betweenthe measured phase shift Δ_(F) and the corresponding value of theenvironmental factor.

In the described embodiments, the sine coil 7 and the cosine coil 9 arearranged so that their relative contributions to the total magneticfield component perpendicular to the PCB 5 vary in accordance withposition along the measurement direction. In particular, the sine andcosine coils have an alternate twisted loop structure. However, it wouldbe apparent to a person skilled in the art that an enormous variety ofdifferent excitation winding geometries could be employed to formtransmit aerials which achieve the objective of causing the relativeproportions of the first and second transmit signals appearing in theultimately detected combined signal to depend upon the position of thesensor element in the measurement direction.

While in the described embodiments, the excitation windings are formedby conductive tracks on a printed circuit board, they could also beprovided on a different planar substrate or, if sufficiently rigid,could even be free standing. Further, it is not essential that theexcitation windings are planar because, for example, cylindricalwindings could also be used with the sensor element moving along thecylindrical axis of the cylindrical winding.

If the inductive sensor is used to measure only an environmental factorsuch as temperature or humidity, only one transmit aerial could be usedas there is no requirement for the phase of the magnetic field to varywith position.

In the first embodiment, the modulating signals are described as digitalrepresentations of sinusoidal signals. This is not strictly necessaryand it is often convenient to use modulating signals that can be moreeasily generated using simple electronics. For example, the modulatingsignals could be digital representations of triangular waveforms. Thephase of the modulation can be decoded in the usual way by only lookingat the fundamental frequency of the modulated signals, i.e. by filteringout the higher harmonics present in the triangular waveform. Note thatsome filtering will be performed as a result of the physical andelectrical properties of, and the electromagnetic coupling between, thetransmit and receive aerials. Alternatively, if no filtering is used,the zero crossing point of the demodulated waveform will still vary withthe target position in some predictable, albeit non-linear, manner whichcould be converted to a linear measurement of position by using look-uptable or a similar technique.

In the first and second embodiments, a quadrature pair of modulationsignals are applied to carrier signals to generate first and secondexcitation signals which are applied to the sine coil 7 and cosine coil9 respectively. However, the use of a quadrature pair of modulationsignals is not essential because it is merely required that theinformation carrying components of the excitation signals are distinctin some way so that the relative contributions from the first and secondexcitation signals can be derived by processing the combined signal. Forexample, the modulation signals could have the same frequency and aphase which differs by an amount other than 90 degrees. Alternatively,the modulation signals could have slightly different frequencies thusgiving rise to a continuously varying phase difference between the twosignals.

In the above described embodiment, a passive resonator is used. However,in some circumstances it may be advantageous to use a powered resonatorso that the signal induced in the resonator is considerably amplified,thus reducing the requirements on the signal processing circuitry.

Instead of detecting the phase of the information carrying components ofthe combined signal, it is also possible to perform parallel synchronousdetection of the combined signal, one synchronous detection using anin-phase modulation signal and the other synchronous detection using aquadrature modulation signal, and then to perform an arctangentoperation on the ratio of the detected magnitudes of the demodulatedsignals. In such an embodiment, by using excitation signals whichcomprise a carrier frequency signal and a modulation signal so that themodulation signals can have a relatively low frequency, the detection ofthe magnitude of the modulation signals and the ensuing arctangentcalculation (or reference to a look-up table) can be performed in thedigital domain after down-conversion from the carrier frequency. Analternative method of detection of the information carrying part of thesignal after down-conversion from the carrier frequency signal tobaseband would be to perform a fast Fourier transform detection. As willbe appreciated, this could be done either using some additionalspecialised dedicated hardware (e.g. an application specific integratedcircuit) or by suitably programming the microprocessor. Such a method ofdetection would be particularly convenient in an arrangement in whichmore than one degree of freedom of movement of a target is to bedetected.

Although synchronous detection is preferred because the phase sensitivenature of the synchronous detection removes noise, alternatively afiltering arrangement could be used to isolate the signals at eachcarrier frequency. For example, the sense signal induced in the sensecoil could be input to a parallel arrangement of bandpass filters, witheach bandpass filter centred at a respective different carrierfrequency. The signal strengths at each frequency can then be comparedin order to determine the noise component at the carrier frequency closeto the resonant frequency f_(res).

In the above described embodiment, the measurement path extends onlyover a single period of the spatial variation of the two transmit coils(i.e. the sine coil 7 and the cosine coil 9). However, this need not bethe case and the measurement path could extend over more or less than asingle period of the spatial variation of the transmit coils. In such acase, it is preferable to include a mechanism for resolving periodambiguity (i.e. the fact that the basic phase of the informationcarrying component of the combined signal will be identical for the samecorresponding position in different spatial periods of the transmitcoils). Mechanisms for overcoming spatial period ambiguity which can beemployed include providing a single reference position detected, forexample, by a single location position sensor (e.g. by having a singlelocalised transmit coil transmitting a third transmit signal at adifferent modulation frequency to add with the first and second transmitaerials, or by using an opto-switch) and thereafter counting the periodsfrom the reference position, and maintaining a record in a registerwithin the microprocessor of the particular period within which thesensor element is currently located. Alternatively, an additional set oftransmit coils transmitting at a different modulation frequency (ortransmitting in a time multiplexed manner), could be used with either aslightly varying spatial frequency to provide a Vernier scale effect, orwith a widely varying spatial frequency to provide coarse positiondetection using a large scale set of transmit coils and fine scaleposition detection using small scale transmission coils.

In the described embodiment, a modulation frequency of 2.5 kHz is usedbecause it is well suited to digital processing techniques. Thisgenerally applies to frequencies in the range 100 Hz to 100 kHz.Preferably, frequencies in the range of 1–10 kHz are used, for example3.9 kHz or 5 kHz.

Although in the first embodiment the PWM type pattern generator isclocked at 2 MHz, other clocking frequencies could be used. Further, theclocking frequency need not be equal to one of the carrier frequencies.

In the described embodiment, a carrier frequency of 2 MHz is used. Usinga carrier frequency above 1 MHz facilitates making the sensor elementsmall. However, in some applications it may be desirable to use acarrier frequency below 100 kHz, for example if a sheet of non-magneticstainless steel separates the sensor element from the excitation andsensor windings, because the skin depth of the non-magnetic stainlesssteel is greater at lower frequencies.

1. An inductive sensor for sensing a parameter, the inductive sensorcomprising: a magnetic field generator operable to generate a magneticfield; a receive aerial electromagnetically coupled to the magneticfield generator via a resonant circuit, said electromagnetic couplingbeing variable in dependence on the sensed parameter so that a sensesignal induced in the receive aerial in response to a magnetic fieldgenerated by the magnetic field generator is representative of thesensed parameter, wherein the magnetic field generator is operable togenerate a magnetic field comprising a first magnetic field component ata first frequency which is operable to induce resonance in the resonantcircuit and a second magnetic field component at a second frequencywhich is not operable to induce resonance in the resonant circuit; and asignal processor operable to process the sense signal to generate asignal component corresponding to a first component of the sense signalat said first frequency adjusted using a second component of the sensesignal at the second frequency, and to determine a value representativeof the sensed parameter using the signal component.
 2. An inductivesensor according to claim 1, wherein the signal processor comprises: afirst synchronous detector operable to detect the first component of thesense signal; a second synchronous detector operable to detect thesecond component of the sense signal; and a combiner operable to combinethe first and second components of the sense signal to generate thesignal component.
 3. An inductive sensor according to claim 1, whereinthe signal processor comprises: a reference signal generator operable togenerate a reference signal having frequency components at the first andsecond frequencies; and a synchronous detector operable to detect thesignal component using the reference signal generated by the referencesignal generator.
 4. An inductive sensor according to claim 1, whereinthe magnetic field generator is operable to generate a magnetic fieldincluding magnetic field components at a plurality of differentfrequencies which are not operable to induce resonance in the resonantcircuit, wherein the signal processor is operable to generate the signalcomponent using components of the sense signal at said plurality ofdifferent frequencies which do not induce resonance.
 5. An inductivesensor according to claim 4, wherein the signal processor comprises: aninterpolator operable to interpolate a noise component at the firstfrequency using the components of the sense signal at said plurality ofdifferent frequencies which do not induce resonance; and a signaladjuster operable to adjust said first component of the sense signal inaccordance with the interpolated noise component.
 6. An inductive sensoraccording to claim 1, wherein the magnetic field generator comprises: atransmit aerial; and a signal generator operable to apply an excitationsignal to the transmit aerial in order to generate said magnetic field.7. An inductive sensor according to claim 6, wherein the transmit aerialcomprises conductive track on a planar substrate.
 8. An inductive sensoraccording to claim 7, wherein the planar substrate is a printed circuitboard.
 9. An inductive sensor according to claim 6, wherein the transmitaerial comprises first and second excitation windings and the receiveaerial comprises a sensor winding, wherein the electromagnetic couplingbetween the first and second excitation windings and the sensor windingvaries along a path in accordance with respective different functions.10. An inductive sensor according to claim 9, wherein the signalprocessor comprises a phase detector operable to measure the phase ofsaid signal component.
 11. An inductive sensor according to claim 1,wherein said magnetic field generator is operable to induce a resonantsignal in the resonant circuit, which in turn is operable to induce anelectromotive force in the receive aerial.
 12. An inductive sensoraccording to claim 6, wherein the transmit aerial and the receive aerialare fixed relative to a first member and the resonant circuit is fixedrelative to a second member, wherein at least one of the first andsecond members is movable relative to the other of the first and secondmembers, and wherein the electromagnetic coupling between the transmitaerial and the receive aerial varies in response to relative movementbetween the first and second members, and wherein the signal processoris operable to determine a value representative of the relative positionof the first and second members.
 13. An inductive sensor according toclaim 1, wherein the receive aerial comprises the resonant circuit. 14.An inductive sensor according to claim 6, wherein the transmit aerial isfixed relative to a first member and the receive aerial is fixedrelative to a second member, wherein at least one of the first andsecond members is movable relative to the other of the first and secondmembers, and wherein the electromagnetic coupling between the transmitaerial and the receive aerial varies in response to relative movementbetween the first and second members, and wherein the signal processoris operable to determine a value representative of the relative positionof the first and second members.
 15. An inductive sensor for sensing aparameter, the inductive sensor comprising: a receive aerialelectromagnetically coupled to a magnetic field generator via a resonantcircuit, said electromagnetic coupling being variable in dependence onthe sensed parameter so that a sense signal induced in the receiveaerial in response to a magnetic field generated by the magnetic fieldgenerator is representative of the sensed parameter, wherein themagnetic field generator is operable to generate a magnetic fieldcomprising a first magnetic field component at a first frequency whichis operable to induce resonance in the resonant circuit and a secondmagnetic field component at a second frequency which is not operable toinduce resonance in the resonant circuit; and a signal processoroperable to process the sense signal to generate a signal componentcorresponding to a first component of the sense signal at said firstfrequency adjusted using a second component of the sense signal at thesecond frequency to reduce a noise component of the first component ofthe sense signal, and to determine a value representative of the sensedparameter using the signal component.
 16. An inductive sensing methodfor sensing a parameter, the method comprising the steps of: generatinga magnetic field comprising a first magnetic field component at a firstfrequency which is operable to induce resonance in a resonant circuitand a second magnetic field component at a second frequency which is notoperable to induce resonance in the resonant circuit; and processing asense signal induced in a receive aerial in response to a magnetic fieldgenerated by the magnetic field generator, wherein the receive aerial iselectromagnetically coupled to the magnetic field via the resonantcircuit with said electromagnetic coupling varying in dependence on thesensed parameter so that the sense signal is representative of thesensed parameter, wherein said processing step comprises generating asignal component corresponding to a first component of the sense signalat said first frequency adjusted using a second component of the sensesignal at the second frequency, and determining a value representativeof the sensed parameter using the signal component.